Method and device for carrier-frequency synchronisation of a vestigial-sideband-modulated signal

ABSTRACT

A method and a device for carrier-frequency synchronisation of a vestigial-sideband-modulated received signal (r VSB (t)) with a carrier signal (e j(2π(f     T     +Δf)t+Δφ) ), which is affected by a frequency offset and/or phase offset (Δf, Δφ), estimates the frequency offset and/or phase offset (Δf, Δφ) of the carrier signal (e j(2π(f     T     +Δf)t+Δφ) ) by means of a maximum-likelihood estimation. For this purpose, the vestigial-sideband-modulated received signal (r VSB (t)) is converted into a modified vestigial-sideband-modulated received signal (x VSB ′(t′)), which provides time-discrete complex rotating phasors (|x VSB (t′)·e j2πΔf·t′+Δφ) , in which only the time-discrete phases (2πΔft′+Δφ) are dependent upon the frequency offset and/or phase offset (Δf, Δφ).

The invention relates to a method and a device for carrier-frequency synchronisation with a vestigial-sideband-modulated signal (VSB).

If transmitters and receivers are synchronised with one another within a transmission system, a transmitter-end adaptation and a receiver-end adaptation of the clock and carrier signal takes place with regard to phase position and frequency respectively. The carrier-frequency synchronisation to be considered below presupposes a received signal, which is synchronised with regard to the clock signal.

Document DE 103 09 262 A1 describes a method for carrier-frequency synchronisation of a signal with digital symbol sequences, in which the frequency offset and/or phase offset of the carrier signal is estimated from the demodulated received signal by means of maximum-likelihood estimation. The received signal containing the digital symbol sequences consists of complex rotating phasors associated with the individual sampling times, of which the discrete phases depend only on the sought frequency offset and/or phase offset of the carrier signal, and of which the discrete amplitudes depend only on the digital symbol values of the demodulated received signal. The maximum-likelihood estimation of the sought frequency offset and/or phase offset of the carrier signal takes place by maximising the likelihood function, which is formed from the sum of the real components of all of the time-discrete, complex rotating phasors of the received signal. Maximising the likelihood function is achieved by rotating the associated, complex rotating phasor of the received signal in a clockwise direction for each sampling time at the level of the sought frequency offset and/or phase offset, so that it is disposed on the real axis. In this manner, it is possible, to obtain the sought frequency offset and/or phase offset of the carrier signal by observing the extreme-values of the likelihood function separately for the frequency offset and/or phase offset.

The time-discrete received signal in DE 103 09 262 A1 provides one complex rotating phasor for each sampling time, of which the phase value depends only upon the frequency offset and/or phase offset of the carrier signal, and of which the amplitude value depends on the symbol value of the received signal sequence to be transmitted at the respective sampling time. A time-discrete received signal of this kind is based upon a comparatively simple modulation, for example, a conventional amplitude modulation. By contrast, a VSB received signal provides no time-discrete, complex rotating phasors, of which the time-discrete phases are dependent only upon the frequency offset and phase offset. In this case, the use of a maximum-likelihood estimation to estimate the sought frequency offset and/or phase offset of the carrier signal in the sense of the method and the device known from DE 103 09 262 A1 therefore fails to achieve the goal.

The invention is accordingly based upon the object of providing a method and a device for estimating the frequency offset and/or phase offset in the carrier signal in the case of a vestigial-sideband-modulated received signal using a maximum-likelihood estimation.

The object of the invention is achieved by a method for carrier-frequency synchronisation with the features of claim 1 and by a device for carrier-frequency synchronisation with the features of claim 10. Further developments of the invention are specified in the dependent claims.

In a first stage, the VSB received signal is converted into a modified VSB signal, which, in an equivalent manner to a quadrature-modulated signal—for example, a PAM, QPSK or π/4-QPSK signal—provides time-discrete, complex rotating phasors consisting respectively of an in-phase component and a quadrature component. The symbol duration of the VSB received signal according to the invention is therefore adjusted to the level of half the symbol duration of a quadrature-modulated signal, and the accordingly-adapted VSB received signal is converted by down mixing into a modified VSB signal, which consists of a complex rotating phasor typical for a quadrature-modulated signal and provides a signal display equivalent to that of an offset QPSK signal.

In a second stage, this modified VSB received signal, which is equivalent to an offset QPSK signal, is additionally converted, in order to realise time-discrete complex rotating phasors, which are dependent only upon the frequency offset and phase offset of the carrier signal. The modified VSB signal is therefore converted by sampling with an oversampling factor of typically eight, estimation filtering with a signal-adapted estimation filter and three further signal-processing stages according to the invention, in order to realise an additionally-modified VSB received signal, of which the time-discrete complex rotating phasors each provide phases, which depend only upon the frequency offset and/or phase offset of the carrier signal used.

The first signal-processing stage involves a further sampling, which generates a time-discrete, modified VSB received signal with two sampling values per symbol period. This accordingly re-sampled VSB received signal contains in each of its discrete, complex rotating phasors an additional phase dependent upon the respective sampling time, which, in the subsequent, second signal-processing stage, is compensated by a respective, inverse phase, in the context of a complex multiplication with a complex rotating phasor. In a third signal-processing stage, the VSB received signal, freed from its additional phase in each of the time-discrete complex rotating phasors, is finally subjected to modulus-scaled squaring, in order to ensure that the amplitude of each time-discrete complex rotating phasor of the modified VSB received signal has a positive value.

With the method according to the invention and the device according to the invention for carrier-frequency synchronisation, a modified VSB received signal, of which the time-discrete complex rotating phasors each provide phases, which are dependent only upon the frequency offset and/or phase offset of the carrier signal used, is therefore formed from the VSB received signal.

The time-discrete phases of the multi-modified, time-discrete VSB received signal are then determined via an argument function, and a phase characteristic is formed.

This phase characteristic of the modified VSB received signal, which is periodic over the period 2·π and non-steady, is then “steadied” at the non-steady points to provide a phase-continuous phase characteristic of the modified VSB received signal.

A phase-continuous phase characteristic of a multi-modified VSB received signal generated in this manner can be subjected to a maximum-likelihood estimation in the sense of DE 103 09 262, in order to determine a frequency offset and/or phase offset possibly occurring in the carrier signal used, thereby approaching the goal of a subsequent carrier-frequency synchronisation of the VSB received signal.

A preferred exemplary embodiment of the method according to the invention for carrier-frequency synchronisation of a VSB signal and the device according to the invention for carrier-frequency synchronisation of a VSB signal are explained in greater detail below with reference to the drawings. The drawings are as follows:

FIG. 1 shows an extended block circuit diagram of the transmission system;

FIG. 2 shows a reduced block circuit diagram of the transmission system;

FIG. 3 shows a block circuit diagram of the device for carrier-frequency synchronisation according to the invention;

FIG. 4 shows a complex phasor diagram of a modified VSB received signal according to the invention;

FIG. 5 shows a flow chart of the method according to the invention for carrier-frequency synchronisation.

Before describing an embodiment of the method according to the invention and the device according to the invention for carrier-frequency synchronisation with a VSB received signal in greater detail with reference to FIGS. 3 to 5, the following paragraphs present a derivation of the required mathematical background.

Accordingly, in a first stage, the VSB signal s_(VSB)(t) is converted according to equation (1) into a modified VSB received signal, which provides a signal display equivalent to an offset QPSK signal with a complex rotating phasor. $\begin{matrix} {{s_{VSB}(t)} = {\sum\limits_{v = {- \infty}}^{+ \infty}{{q(v)} \cdot {\delta\left( {t - {v \cdot T_{VSB}}} \right)}}}} & (1) \end{matrix}$ In this equation, the values q(v) in a 2VSB signal represent the symbol sequence with the symbol alphabet {+1+pilot, −1+pilot} and the symbol duration T_(VSB). A pilot carrier, for which the condition pilot=0 applies, is conventionally contained therein. In the deliberations presented below, ideal conditions—pilot=0—are assumed.

In a complex baseband model of a transmission system 1 for time-continuous complex signals, as shown in FIG. 1, the VSB signal S_(VSB)(t) is supplied to a transmission filter 2. As in the case of a quadrature-modulated signal—such as a PAM, QPSK or n/4-QPSK modulated signal—, this is realised as a cosine filter with a transmission function H_(S)(f) and a roll-off factor r, as shown in equation (2). $\begin{matrix} {{H_{s}(f)} = \left\{ {\begin{matrix} {1} & {{für}{{f{{< \frac{f_{s}}{2}}}}}} \\ {\cos\quad\left\lbrack {\frac{\pi\quad{{f}}}{2{r \cdot f_{s}}} - \frac{\pi\quad\left( {1 - r} \right)}{4r}} \right\rbrack} & {{{{für}\left( {1 - r} \right)}\frac{f_{s}}{2}} < {{f{{\leq {\left( {1 + r} \right)\frac{f_{s}}{2}}}}}}} \\ {0} & {{{{für}\left( {1 + r} \right)}\frac{f_{s}}{2}} < {{f}}} \end{matrix}\left\lbrack {{für} = {for}} \right\rbrack} \right.} & (2) \end{matrix}$ By contrast with the transmission filter in a PAM, QPSK or π/4 QPSK modulated signal, the transmission filter 2 associated with a VSB signal S_(VSB)(t) is a cosine filter symmetrical to the frequency ${f = {\frac{1}{4} \cdot f_{SVSB}}},$ wherein f_(SVSB) is the symbol frequency of the VSB signal inverse to the symbol period T_(VSB). Its transmission function H_(SVSB)(f) is therefore derived from a displacement of the transmission function H_(S)(f) of a quadrature-modulated signal by the frequency $f = {\frac{1}{4} \cdot f_{SVSB}}$ in the sense of equation (3). $\begin{matrix} {{H_{SVSB}(f)} = {H_{s}\left( {f - {\frac{1}{4} \cdot f_{SVSB}}} \right)}} & (3) \end{matrix}$ The context shown in equation (4) applies for the symbol rate f_(s) from equation (2), which relates to classical, quadrature-modulated signals, and for the symbol frequency of a VSB signal f_(SVSB): $\begin{matrix} {f_{s} = {\frac{1}{2} \cdot f_{SVSB}}} & (4) \end{matrix}$ The impulse response h_(SVSB)(t) of the transmission filter 2 for a VSB signal is therefore derived from the following equation (5): $\begin{matrix} {{h_{SVSB}(t)} = {{h_{s}(t)} \cdot {\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t}}} & (5) \end{matrix}$ The VSB signal S_(FVSB)(t) disposed at the output of the transmission filter 2 is derived from a convolution of the VSB received signal S_(VSB)(t) according to equation (1) with the impulse response h_(SVSB)(t) of the transmission filter 2 according to equation (5) and is described mathematically by equation (6), which is mathematically converted over several stages: $\begin{matrix} \begin{matrix} {{s_{FVSB}(t)} = {\left( {{h_{s}(t)} \cdot {\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t}} \right) \star {\sum\limits_{v = {- \infty}}^{+ \infty}{{q(v)} \cdot {\delta\left( {t - {v \cdot T_{VSB}}} \right)}}}}} \\ {= {\sum\limits_{v = {- \infty}}^{+ \infty}{{q(v)} \cdot {h_{s}\left( {t - {v \cdot T_{VSB}}} \right)} \cdot {\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}{({t - {v \cdot T_{VSB}}})}}}}} \\ {= {{\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t} \cdot {\sum\limits_{v = {- \infty}}^{+ \infty}{{q(v)} \cdot {\mathbb{e}}^{{- j}\frac{\quad\pi}{2}v} \cdot {h_{s}\left( {t - {v \cdot T_{VSB}}} \right)}}}}} \end{matrix} & (6) \end{matrix}$ According to equation (7), the value b(v) is introduced for the term ${q(v)} \cdot {\mathbb{e}}^{{- j}\frac{\quad\pi}{2}v}$ in equation (6): $\begin{matrix} {{b(v)}\text{:}{= {q(v)} \cdot {\mathbb{e}}^{{- j}\frac{\quad\pi}{2}v}}} & (7) \end{matrix}$ If the value b(v) is observed separately for even-numbered and odd-numbered v, then for even-numbered v=2n (n: integer), the mathematical relationship of equation (8) is obtained, which, after mathematical conversion, gives real values a_(R)(n): $\begin{matrix} \begin{matrix} {{b(v)}{_{v = {2n}}{= {{\mathbb{e}}^{{- j}\frac{\pi}{2}2n} \cdot {q\left( {2n} \right)}}}}} \\ {= {{\mathbb{e}}^{{- j}\quad\pi\quad n} \cdot {q\left( {2n} \right)}}} \\ {= {{\left( {- 1} \right)^{n} \cdot {q\left( {2n} \right)}}\text{:} = {a_{R}(n)}}} \end{matrix} & (8) \end{matrix}$ With odd-numbered v=2n+1 (n: integer), for the value b(v), the mathematical relationship of equation (9), is obtained, which, after mathematical conversion, gives complex values j·a_(x)(n): $\begin{matrix} \begin{matrix} {{b(v)}{_{v = {{2n} + 1}}{= {{\mathbb{e}}^{{- j}\frac{\pi}{2}{({{2n} + 1})}} \cdot {q\left( {{2n} + 1} \right)}}}}} \\ {= {{j \cdot \left( {- 1} \right)^{n + 1} \cdot {q\left( {{2n} + 1} \right)}}\text{:}}} \\ {= {j \cdot {a_{I}(n)}}} \end{matrix} & (9) \end{matrix}$ The term ${q(v)} \cdot {\mathbb{e}}^{{- j}\frac{\pi}{2}v}$ in equation (6) can be presented according to equation (10) for even-numbered v=2n as the even-numbered term b(v)|_(v=2n), and for odd-numbered v=2n+1, as the odd-numbered term b(v)|_(v=2n+1): $\begin{matrix} {{{q(v)} \cdot {\mathbb{e}}^{{- j}\frac{\pi}{2}v}} = {{b(v)}{_{v = {2n}}{+ {b(v)}}}_{v = {{2n} + 1}}}} & (10) \end{matrix}$ The mathematical relationship for the output signal S_(FVSB)(t) at the output of the transmission filter 2 in equation (6) can therefore be converted according to equation (10), taking into consideration equation (8) and (9), as shown in equation (11): $\begin{matrix} {{s_{FVSB}(t)} = {{\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t} \cdot \left( {{\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{S}\left( {t - {2{n \cdot T_{VSB}}}} \right)}}} + {j \cdot {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{S}\left( {t - {2{n \cdot T_{VSB}}} - T_{{{VSB})})}} \right.}}}}} \right.}} & (11) \end{matrix}$ The subsequent lag element 3 models the time offset ε·T occurring as a result of the absent or inadequate synchronisation between the transmitter and the receiver, which is derived from a timing offset ε. The timing offset ε in this context can adopt positive and negative values typically between ±0.5. The filtered symbol sequence S_(εVSB)(t) at the output of the lag element 3, which takes the time offset ε·T into consideration, is therefore derived according to equation (12): $\begin{matrix} {{s_{ɛ\quad{VSB}}(t)} = {{\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t} \cdot \left( {{\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{S}\left( {t - {ɛ \cdot T_{VSB}} - {2{n \cdot T_{VSB}}}} \right)}}} + {j \cdot {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{S}\left( {t - {ɛ \cdot T_{VSB}} - {2{n \cdot T_{VSB}}} - T_{VSB}} \right)}}}}} \right)}} & (12) \end{matrix}$ The lag-affected, filtered symbol sequence S_(εVSB)(t) is mixed in a VSB modulator—modelled as a multiplier 4 in FIG. 1—with a complex carrier signal e^(j(2π(f) ^(T) ^(+Δf)t+Δφ)) to form a VSB-modulated HF-transmission signal S_(HFVSB)(t). The carrier signal e^(j(2π(f) ^(T) ^(+Δf)t+Δφ)) has a carrier frequency f_(T), which has a frequency offset Δf and phase offset Δφ. The mathematical context for the VSB-modulated HF transmission signal S_(HFVSB)(t) is presented in equation (13): $\begin{matrix} {{s_{HFVSB}(t)} = {{\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t} \cdot \left\lbrack {{\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{S}\left( {t - {ɛ\quad T_{VSB}} - {2{nT}_{VSB}}} \right)}}} + {j \cdot {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{S}\left( {t - {ɛ\quad T_{VSB}} - {2{nT}_{VSB}} - T_{VSB}} \right)}}}}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad{\pi{({f_{T} + {\Delta\quad f}})}}t} + {\Delta\quad\varphi}})}}}} & (13) \end{matrix}$ Additive, white Gaussian noise (AWGN) n(t), which provides a real and an imaginary component n_(R)(t) and n_(I)(t) as shown in equation (14), is superimposed additively onto the VSB-modulated transmission signal S_(HFVSB)(t) on the transmission path between the transmitter and a receiver. n(t)=n _(R)(t)+j·n _(I)(t)  (14) The VSB-modulated HF received signal r_(HFVSB)(t) arriving in the receiver is therefore obtained from equation (15): r _(HFVSB)(t)=s _(HFVSB)(t)+n(t)  (15) In the receiver, the VSB-modulated received signal r_(HFVSB)(t) with superimposed noise n(t) is mixed down into the baseband with the carrier signal e^(−j2πf) ^(T) ^(t) in a demodulator—modelled as the multiplier 5 in FIG. 1. The demodulated VSB received signal r_(VSB)(t) at the output of the demodulator 5 is derived according to the equation (16): $\begin{matrix} {{r_{VSB}(t)} = {{{{s_{ɛ\quad{VSB}}(t)} \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad f} + {\Delta\quad\varphi}})}}} + {n(t)}} = {{{\mathbb{e}}^{j\frac{2\quad\pi}{4T_{VSB}}t} \cdot \left\lbrack {{\sum\limits_{n = {- \infty}}^{+ \infty}{{{a_{R}(n)} \cdot h_{S}}\left( {t - {ɛ\quad T_{VSB}} - {nT}_{VSB}} \right)}} + {j \cdot {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{S}\left( {t - {ɛ\quad T_{VSB}} - {2{nT}_{VSB}} - T_{VSB}} \right)}}}}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{ft}} + {\Delta\quad\varphi}})}}} + {n(t)}}}} & (16) \end{matrix}$ As can be seen from equation (16), some of the system-theoretical effects of the modulator 4 and the demodulator 6 in the transmission system 1 on the VSB-modulated signal are cancelled, so that the modulator 5 and the demodulator 6 in FIG. 1 can be replaced by a single multiplier 7 as shown in the reduced block circuit diagram in FIG. 2.

If the VSB baseband received signal r_(VSB)(t) according to equation (16) is mixed with a signal ${\mathbb{e}}^{{- j}\frac{2\quad\pi}{4T_{VSB}}t},$ if the symbol duration T_(VSB) of the VSB signal according to the equation (17) is adjusted to half the symbol duration T_(S) of a quadrature-modulated signal and if the frequency-displaced cosine filter $H_{S}\left( {f - {\frac{1}{4} \cdot f_{SVSB}}} \right)$ according to equation (2) of a quadrature-modulated signal is used as the transmission filter of the VSB signal, the mathematical relationship of equation (18) is derived, starting from equation (16), for the modified VSB baseband received signal r_(VSB)′(t). $\begin{matrix} {T_{VSB} = {\frac{1}{2} \cdot T_{S}}} & (17) \\ {{r_{VSB}^{\prime}(t)} = {{\left( {{\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{S}\left( {t - {ɛ\quad T_{S}} - {n \cdot T_{S}}} \right)}}} + {j \cdot {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{S}\left( {t - {ɛ\quad T_{S}} - {n \cdot T_{S}} - \frac{T_{S}}{2}} \right)}}}}} \right) \cdot {\mathbb{e}}^{{j{({{\Delta\quad\text{ft}} + {\Delta\varphi}})}}\quad}} + {n(t)}}} & (18) \end{matrix}$ The mathematical term for the modified VSB baseband received signal r_(VSB)′(t) in equation (18) corresponds to the signal display for an offset QPSK signal, of which the quadrature component is phase-displaced relative to the in-phase component by half of one symbol length T_(S).

The modified VSB baseband received signal r_(VSB)′(t) is supplied to a sampling and holding element 7, referred to below as the second sampling and holding element for an oversampling of the filtered, modified VSB baseband received signal at a sampling rate f_(A), which is increased by comparison with the symbol frequency f_(SVSB) of the received signal r_(VSB)′(t) by the oversampling factor os. In this context, the oversampling factor os has a value of 8, as shown in detail in [1]: K. Schmidt: “Digital clock recovery for bandwidth-efficient mobile telephone systems” [Digitale Taktrückgewinnung für bandbreiteneffiziente Mobilfunksysteme], 1994, ISBN 3-18-14 7510-6.

After the sampling of the modified VSB baseband received signal r_(VSB)′(t), an estimation filtering of the signal takes place in an estimation filter 8. The estimation filter 8 has the task of minimising data-dependent jitter in the signal. If the transmission filter 2 according to equation (2) has a frequency spectrum H_(S)(f), which corresponds to a cosine filter with a roll-off factor r, the frequency spectrum H_(EST)(f) of the estimation filter 8 must be designed according to equation (19) dependent upon the frequency spectrum H_(S)(f) of the transmission filter 2 in order to minimise data-dependent jitter in the modified VSB baseband received signal r_(VSB)′(t), as shown in [1]. $\begin{matrix} {{H_{EST}(f)} = \left\{ {\begin{matrix} {{H_{S}\left( {f - f_{S}} \right)} + {H_{S}\left( {f + f_{S}} \right)}} & {{für}{{f{{\leq {\frac{f_{S}}{2}\left( {1 + r} \right)}}}}}} \\ {beliebig} & {{{für}\frac{f_{S}}{2}\left( {1 + r} \right)} < {f} \leq f_{S}} \\ {0} & {{{für}\quad f_{S}} < {f}} \end{matrix}\left\lbrack {{{beliebig} = {random}};{{für} = {for}}} \right\rbrack} \right.} & (19) \end{matrix}$ The frequency response H_(GES)(f)=H_(S)(f)·H_(EST)(f) of the transmission system as a whole, consisting of transmission filter 2 and estimation filter 8, can be interpreted according to the equation (20) as a low pass filter H_(GES0)(f) symmetrical to the frequency f=0 with a bandwidth of ${\frac{f_{S}}{2} \cdot r},$ which is frequency-displaced in each case by ${\pm \frac{f_{S}}{2}}\text{:}$ $\begin{matrix} \begin{matrix} {{H_{GES}(f)} = {{H_{{GES}\quad 0}(f)} \star \left( {\delta\left( {f - \frac{f_{S}}{2} + {\delta\left( {f + \frac{f_{S}}{2}} \right)}} \right)} \right.}} \\ {= {{H_{{GES}\quad 0}\left( {f - \frac{f_{S}}{2}} \right)} + {H_{{GES}\quad 0}\left( {f + \frac{f_{S}}{2}} \right)}}} \end{matrix} & (20) \end{matrix}$ The corresponding impulse response h_(GES)(t) is therefore derived according to equation (21): $\begin{matrix} {{h_{GES}(t)} = {{{h_{{GES}\quad 0}(t)} \cdot \left( {{\mathbb{e}}^{{j2}\quad\pi\frac{f_{S}}{2}t} + {\mathbb{e}}^{{- {j2}}\quad\pi\frac{f_{S}}{2}t}} \right)} = {{{h_{{GES}\quad 0}(t)} \cdot \cos}\quad\left( {2\quad\pi\frac{f_{S}}{2}t} \right)}}} & (21) \end{matrix}$ The signal v_(VSB)′(t) at the output of the estimation filter 8 can therefore be obtained according to equation (22), in that, in the modified VSB received signal r_(VSB)′(t) in the baseband, as shown in equation (18), the impulse response h_(S)(t) of the transmission filter is replaced with the impulse response h_(GES)(t) of the transmission system as a whole. $\begin{matrix} {{v_{VSB}^{\prime}(t)} = {{\left\lbrack {{\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{GES}\left( {t - {ɛ\quad T_{S}} - {nT}_{S}} \right)}}} + {j \cdot {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{GES}\left( {t - {ɛ\quad T_{S}} - \frac{T_{S}}{2} - {nT}_{S}} \right)}}}}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{ft}} + {\Delta\quad\varphi}})}}} + {n(t)}}} & (22) \end{matrix}$ Starting from equation (22), the impulse response h_(GES)(t−εT_(S)−nT_(S)) can be described according to equation (23): $\begin{matrix} {{h_{GES}\left( {t - {ɛ\quad T_{S}} - {nT}_{S}} \right)} = {{{h_{{GES}\quad 0}\left( {t - {ɛ\quad T_{S}} - {nT}_{S}} \right)} \cdot \left( {- 1} \right)^{n} \cdot \cos}\quad\left( {2\quad\pi\quad\frac{f_{S}}{2}\left( {t - {ɛ\quad T_{S}}} \right)} \right)}} & (23) \end{matrix}$ Similarly, the mathematical relationship for the impulse response $h_{GES}\left( {t - {ɛ\quad T_{S}} - \frac{T_{S}}{2} - {nT}_{S}} \right)$ can be determined in equation (24). $\begin{matrix} {{h_{GES}\left( {t - {ɛ\quad T_{S}} - \frac{T_{S}}{2} - {nT}_{S}} \right)} = {{{h_{{GES}\quad 0}\left( {t - {ɛ\quad T_{S}} - \frac{T_{S}}{2} - {nT}_{S}} \right)} \cdot \left( {- 1} \right)^{n} \cdot \sin}\quad\left( {2\quad\pi\quad\frac{f_{S}}{2}\left( {t - {ɛ\quad T_{S}}} \right)} \right)}} & (24) \end{matrix}$ On the basis of the mathematical terms in the equations (23) and (24), the combined terms can be formulated as in equations (25) and (26), and accordingly, the mathematical context for the output signal v_(VSB)′(t) of the estimation filter 8 in the case of an excitation of the transmission system 1 with a VSB signal s_(VSB)(t) from equation (22) can be carried over according to equation (27). $\begin{matrix} {{R(t)} = {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{{GES}\quad 0}\left( {t - {ɛ\quad T_{S}} - {nT}_{S}} \right)} \cdot \left( {- 1} \right)^{n}}}} & (25) \\ {{I(t)} = {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{I}(n)} \cdot {h_{{GES}\quad 0}\left( {t - {ɛ\quad T_{S}} - \frac{T_{S}}{2} - {nT}_{S}} \right)} \cdot \left( {- 1} \right)^{n}}}} & (26) \\ {{v_{VSB}^{\prime}(t)} = {\left\lbrack {{{{R(t)} \cdot \cos}\quad\left( {2\quad\pi\quad\frac{f_{S}}{2}\left( {t - {ɛ\quad T_{S}}} \right)} \right)} + {{j \cdot {I(t)} \cdot \sin}\quad\left( {2\quad\pi\quad\frac{f_{S}}{2}\left( {t - {ɛ\quad T_{S}}} \right)} \right)}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{ft}} + {\Delta\quad\varphi}})}}}} & (27) \end{matrix}$ The signal v_(VSB)′(t) at the output of the estimation filter 8 according to equation (27) is delayed in a subsequent time-delay unit 9 by the timing offset −{circumflex over (ε)}·T_(S). The estimated timing offset {circumflex over (ε)}, which is determined by an estimation unit, not illustrated here, for estimating the timing offset {circumflex over (ε)} of a VSB-modulated signal, corresponds, in the case of an optimum clock synchronisation, to the actual timing offset ε of the VSB modulated signal v_(VSB)′(t). In this case, the output signal v_(εVSB)′(t) of the time-delay unit 9 according to equation (28) is completely freed from its timing offset. $\begin{matrix} {{v_{ɛ\quad{VSB}}^{\prime}(t)} = {\left\lbrack {{{{R_{ɛ}(t)} \cdot \cos}\quad\left( {2\quad\pi\quad{\frac{f_{S}}{2} \cdot t}} \right)} + {{j \cdot {I_{ɛ}(t)} \cdot \sin}\quad\left( {2\quad\pi\quad{\frac{f_{S}}{2} \cdot t}} \right)}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t}} + {\Delta{\quad\quad}\phi}})}}}} & (28) \end{matrix}$ The associated combined terms R_(ε)(t) and I_(ε)(t) freed from the timing offset ε·T. are derived according to equations (29) and (30): $\begin{matrix} {{R_{ɛ}(t)} = {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{R}(n)} \cdot {h_{{GES}\quad 0}\left( {t - {nT}_{S}} \right)} \cdot \left( {- 1} \right)^{n}}}} & (29) \\ {{I_{ɛ}(t)} = {\sum\limits_{n = {- \infty}}^{+ \infty}{{a_{1}(n)} \cdot {h_{{GES}\quad 0}\left( {t - \frac{T_{S}}{2} - {nT}_{S}} \right)} \cdot \left( {- 1} \right)^{n}}}} & (30) \end{matrix}$ It is evident from equations (28), (29) and (30) that the clock-synchronised, modified VSB baseband received signal v_(εVSB)′(t) does not provide the time-discrete form according to equation (31) required in order to use the maximum-likelihood method to determine the frequency-offset and phase-offset estimate Δ{circumflex over (f)} and Δ{circumflex over (φ)}: r(t′)=|r(t′)·e ^(j(2πΔft′+Δφ))  (31) According to the invention, the clock-synchronised, modified VSB baseband received signal v_(εVSB)′(t) is therefore converted, as will be shown below, into a form corresponding to equation (31).

For this purpose, if the output signal v_(VSB)′(t) of the time-delay unit 9 is observed only at the discrete timing points ${t^{\prime} = {{\mu \cdot \frac{T_{S}}{2}}\left( {{\mu = 0},1,2,{{\ldots\quad{2 \cdot N}} - 1}} \right)}},$ then the output signal v_(εVSB)′(t′) of the time-delay unit 9 is composed according to equations (32a), (32b), (32c), and (32d) dependent upon the observed timing point, only of a purely real or purely imaginary component and a complex rotating phasor e^(j(2πΔf·t+Δφ)): $\begin{matrix} {{t^{\prime} = {{{0 \cdot \frac{T_{s}}{2}}\text{:}{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)}} = {\left\lbrack {R_{ɛ}\left( t^{\prime} \right)} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}}{{v_{ɛ\quad{VSB}}^{\prime}(t)} = {{R_{ɛ}(t)} \cdot {\mathbb{e}}^{j{({0\frac{\pi}{2}})}} \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t}} + {\Delta\quad\phi}})}}}}} & \left( {32a} \right) \\ {{t^{\prime} = {{{1 \cdot \frac{T_{s}}{2}}\text{:}{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)}} = {\left\lbrack {j \cdot {I_{ɛ}\left( t^{\prime} \right)}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}}{{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)} = {{I_{ɛ}\left( t^{\prime} \right)} \cdot {\mathbb{e}}^{j{({1\frac{\pi}{2}})}} \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}} & \left( {32b} \right) \\ {{t^{\prime} = {{{2 \cdot \frac{T_{s}}{2}}\text{:}{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)}} = {\left\lbrack {- {R_{ɛ}\left( t^{\prime} \right)}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}}{{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)} = {{R_{ɛ}\left( t^{\prime} \right)} \cdot {\mathbb{e}}^{j{({2\frac{\pi}{2}})}} \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}} & \left( {32c} \right) \\ {{t^{\prime} = {{{3 \cdot \frac{T_{s}}{2}}\text{:}{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)}} = {\left\lbrack {{- j} \cdot {I_{ɛ}\left( t^{\prime} \right)}} \right\rbrack \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}}{{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)} = {{I_{ɛ}\left( t^{\prime} \right)} \cdot {\mathbb{e}}^{j{({3\frac{\pi}{2}})}} \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}} & \left( {32d} \right) \\ \quad & \quad \end{matrix}$ The combined terms R_(ε)(t′) and I_(ε)(t′) according to equations (29) and (30) represent real-value low-pass signals, which can be either positive or negative, because of the statistical distribution of the symbol sequences a_(R)(n) and a_(I)(n).

In the following section, they are described respectively by the time-dependent real-value amplitude A(t′). Accordingly, for the output signal V_(εVSB)′(t′) of the time-delay unit 9 at the individual times $t^{\prime} = {\mu \cdot \frac{T_{S}}{2}}$ μ = 0, 1, 2, …  2 ⋅ N − 1, instead of timing-related individual equations (32a), (32b), (32c) and (32d), a single mathematical equation (33) containing all of the timings is obtained for the output signal v_(εVSB)′(t′) of the time-delay unit 9: $\begin{matrix} {{{v_{ɛ\quad{VSB}}^{\prime}\left( t^{\prime} \right)} = {{A\left( t^{\prime} \right)} \cdot {\mathbb{e}}^{j{({\mu\frac{\pi}{2}})}} \cdot {\mathbb{e}}^{j{({{2\quad\pi\quad\Delta\quad{f \cdot t^{\prime}}} + {\Delta\quad\phi}})}}}}{{{for}\quad t^{\prime}} = {\mu \cdot \frac{T_{S}}{2}}}} & (33) \end{matrix}$ If the time-discrete output signal v_(εVSB)′(t′) of the time-delay unit 9 is frequency-displaced at the individual sampling times ${t^{\prime} = {{{\mu \cdot \frac{T_{S}}{2}}\quad\mu} = 0}},1,2,{{\ldots\quad{2 \cdot N}} - 1}$ by a factor ${- \frac{\pi}{2}},$ a frequency-displaced, time-discrete signal w_(VSB)′(t′) according to equation (34) is obtained from the time-discrete output signal v_(εVSB)′(t′) of the time-delay unit 9, which, by comparison with the time-discrete output signal v_(εVSB)′(t′) of the time-delay unit 9, is freed from the complex term ${\mathbb{e}}^{{j\mu}\frac{\pi}{2}}\text{:}$  w _(VSB)′(t′)=A(t′)·e ^((2πΔf·t′+Δφ))  (34) Since the amplitude A(t′) of the signal w_(VSB)′(t′) can adopt positive and negative values, a modulus for the amplitude must be formed. A modulus for the amplitude A(t′) of a complex signal is formed by squaring and subsequent division by the modulus of the complex signal. The phase of the complex signal is doubled by this process, but the modulus remains unchanged.

The use of a squaring of the signal w_(VSB)′(t′) and subsequent division by the modulus of the signal w_(VSB)′(t′) leads to the signal x_(VSB)′(t′) according to equation (35), which can be interpreted as a time-discrete, complex rotating phasor with a time-discrete amplitude |A(t′)| and a time discrete phase 2·(2πΔft′+Δφ)=2·(ΔωμT_(S)+Δφ) in the sense of FIG. 4 and which has the form according to equation (31) appropriate for a maximum-likelihood estimation of the frequency offset and phase offset of the carrier signal: x _(VSB)′(t′)=|A(t′)|·e ^(j2(2πΔf·t′+Δφ)) +n(t′)  (35) Moreover, in equation (34), the additive interference n(t′) is also taken into consideration, which, in a good approximation, is un-correlated and provides a Gaussian distribution. Accordingly, the optimum estimated value for Δ{circumflex over (f)} and Δ{circumflex over (φ)} is obtained by maximising the maximum-likelihood function L(Δ{circumflex over (f)},Δ{circumflex over (φ)}), which according to equation (36), corresponds to a maximising of the real components of all time-discrete, complex rotating phasors of the signal x_(VSB)′(t′): $\begin{matrix} {{L\left( {{\Delta\quad\hat{f}},{\Delta\quad\hat{\varphi}}} \right)} = {{Re}\left\{ {\sum\limits_{\mu}{{{x_{VSB}^{\prime}\left( {t^{\prime} = {\mu \cdot \frac{T_{s}}{2}}} \right)}} \cdot {\mathbb{e}}^{{- j}\quad 2{({{2\quad\pi\quad\Delta\hat{f}\quad\mu\frac{T_{s}}{2}} + {\Delta\quad\hat{\varphi}}}}}}} \right.}} & (36) \end{matrix}$ Maximising the real components of all time-discrete, complex rotating phasors of the signal x_(VSB)′(t′) can be interpreted as a “turning back” of the time-discrete, complex rotating phasors of the signal x_(VSB)′(t′) respectively by the phase angle $2 \cdot \left( {{2\quad\pi\quad\Delta\quad f\quad\mu\frac{T_{s}}{2}} + {\Delta\quad\varphi}} \right)$ until these coincide with the real axis in the complex plane.

With reference to the derivation of the mathematical background, a description of the device according to the invention for carrier-frequency synchronisation with a VSB-modulated signal according to FIG. 3 and of the method according to the invention for carrier-frequency synchronisation with a VSB-modulated signal according to FIG. 5 is provided below.

In the case of an inverted position of the sideband, the VSB baseband received signal r_(VSB)(t) according to equation (16) is subjected, in a unit for sideband mirroring 10 in the device according to the invention as shown in FIG. 3, to a mirroring of the sideband at the carrier frequency f_(T) into the normal position.

Following this, the VSB baseband received signal r_(VSB)(t) is mixed down in a down mixer 11 by means of a mixer signal ${\mathbb{e}}^{{- j}\frac{2\quad\pi}{4T_{VSB}}t}$ by the frequency $\frac{f_{SVSB}}{4}$ into a modified VSB baseband received signal r_(VSB)′(t) according to equation (18).

A downstream sampling and holding element 7 with an oversampling factor os samples the modified VSB the baseband received signal r_(VSB)′(t). An estimation filtering in the sense of equation (22) or respectively (27) also takes place in a signal-adapted estimation filter 8. A clock synchronisation of the output signal v_(VSB)′(t) of the estimation filter 8 by the timing offset −{circumflex over (ε)}·T_(S) is carried out in a downstream time-delay unit 9 according to equation (28). The estimated timing offset {circumflex over (ε)}, which is determined by an estimation unit, not illustrated here, for the estimation of the timing offset {circumflex over (ε)} of a VSB-modulated received signal, corresponds, in the case of an optimum clock synchronisation, to the actual timing offset ε of the VSB-modulated baseband received signal r_(VSB)′(t)

The clock-synchronised output signal v_(εVSB)′(t) of the time-delay unit 9 is sampled down in a sampling and holding element 12 referred to below as the first sampling and holding element to two sampling values per symbol period T_(S).

The output signal v_(εVSB)′(t′) of the first sampling and holding element 12 is supplied to a complex multiplier 13, in which it is subjected to a sampling-time-related phase offset by the phase angle ${- \mu} \cdot {\frac{T_{S}}{2}.}$

The output signal w_(VSB)′(t′) of the complex multiplier 13, accordingly phase-displaced in its phase relative to the signal v_(εVSB)′(t′), is supplied to a unit for modulus-scaled squaring 14, consisting of a squaring unit, a modulus former and a divider connected downstream of the squaring unit and the modulus former, in which a modulus for its amplitude is formed and its phase is doubled.

The signal at the output of the unit for modulus-scaled squaring 14 represents the modified VSB baseband received signal x_(VSB)′(t′), which the signal-processing unit 15 has generated from the clock-synchronised VSB baseband received signal v_(εVSB)′(t) by undersampling in the first sampling and holding element 12, by phase displacement in the complex multiplier 13 and by modulus formation of the amplitude or respectively doubling of the phase in the unit for modulus-scaled squaring 14.

The estimated values Δ{circumflex over (f)} and Δ{circumflex over (φ)} for the frequency offset and phase offset of the carrier signal are determined, as described, for example, in DE 103 09 262 A1, from the time-discrete, modified VSB a baseband received signal x_(VSB)′(t′) in a subsequent maximum-likelihood estimator 18.

A frequency offset and phase offset estimator, such as that disclosed in DE 103 09 262, which avoids any 2π slips occurring in the phase characteristic—so-called “cycle slips”, which, in the case of a phase regression, result, through small amplitudes of the time-discrete, modified received signal x(t′), from the superimposed interference, can be used as a maximum-likelihood estimator. Accordingly, the phase regression cannot be used for this application.

The method according to the invention for carrier-frequency synchronisation of a VSB-modulated signal is described below with reference to FIG. 5.

In procedural stage S10, the sideband of the VSB baseband received signal r_(VSB)(t) is mirrored by the carrier frequency f_(T) from an inverted position into a normal position, if the sideband is disposed in the inverted position.

In the next procedural stage S20, the VSB baseband received signal r_(VSB)(t) is mixed down with a mixer signal ${\mathbb{e}}^{{- j}\frac{2\quad\pi}{4T_{VSB}}t}$ by the frequency $\frac{f_{SVSB}}{4}$ into a modified baseband received signal r_(VSB)′(t′) according to equation (18).

In the next procedural stage S30, the modified VSB baseband received signal r_(VSB)′(t) is sampled in a second sampling with an oversampling factor of typically eight.

The sampled, modified VSB baseband received signal e_(VSB)′(t′) is supplied, in procedural stage S40, to an estimation filter according to equations (22) and (27) respectively, which minimises data-dependent jitter in the sampled, modified baseband received signal e_(VSB)′(t).

In the next procedural stage S50, a clock-synchronisation of the sampled and filtered, modified VSB baseband received signal v_(εVSB)′(t) takes place according to equation (28) by means of a time-delay unit 9, which receives the estimated timing offset {circumflex over (ε)} from an estimator, which is not described in greater detail here.

In the next procedural stage S60, an additional sampling takes place—a first sampling—of the clock-synchronised VSB baseband received signal v_(εVSB)′(t) at two sampling values per symbol period T_(S) according to equation (33).

In the next procedural stage S70, the additionally sampled clock-synchronised VSB baseband received signal v_(εVSB)′(t′) is frequency displaced by complex multiplication with a sampling-time-related multiplication factor ${\mathbb{e}}^{{- j}\quad\mu\frac{\pi}{2}}$ to compensate the respective inverse complex factor ${\mathbb{e}}^{j\quad\mu\frac{\pi}{2}}$ in the additionally sampled signal v_(εVSB)′(t′) according to equation 34.

The next procedural stage S80 contains the modulus formation of the time-discrete amplitudes A(t′) and a doubling of the time-discrete phases 2πΔft′+Δφ of the frequency-displaced, additionally-sampled and clock-synchronised VSB baseband received signal w_(VSB)′(t′) according to equation (35).

In the next procedural stage S90, the time-discrete, modified VSB baseband received signal x_(VSB)′(t′) obtained from the clock-synchronised VSB baseband received signal v_(εVSB)′(t) in procedural stages S60, S70 and S80 by means of the signal-processing unit 15 is used to determine the frequency offset and phase offset value Δ{circumflex over (f)} and Δ{circumflex over (φ)} of the carrier signal by means of maximum-likelihood estimation according to equation (36). The maximum-likelihood estimator used should ideally be able to overcome any phase slips—so-called “cycle slips” resulting from interference signals, which are superimposed on the modified VSB baseband received signal v_(VSB)′(t) in the case of small amplitudes of the modified VSB baseband received signal v_(εVSB)(t), and is disclosed, for example, in DE 103 09 262 A1. 

1. Method for carrier-frequency synchronisation of a carrier signal (e^(j(2π(f) ^(T) ^(+Δf)t+Δφ))) affected by a frequency offset and/or a phase offset (Δf, Δφ) by estimating the frequency offset and/or phase offset (Δf, Δφ) of the carrier signal (e^(j(2π(f) ^(T) ^(+Δf)t+Δφ))) by means of a maximum-likelihood estimation (S100) from a received signal (r(t′)) with time-discrete, complex rotating phasors (|r(t′)|·e^(j2πΔf·t′+Δφ)), in which only the time-discrete phases (2·(2πΔft′+Δφ)) are dependent upon the frequency offset and/or phase offset (Δf, Δφ), wherein the received signal (r(t′)) is a vestigial-sideband-modulated received signal (r_(VSB)(t)), which is converted for the maximum-likelihood estimation into a modified vestigial-sideband-modulated received signal (x_(VSB)′(t′)) with time-discrete, complex rotating phasors (|A(t′)|·e^(j2(2πΔf·t′+Δφ), in which only the time-discrete phases ()2·(2πΔft′+Δφ)) are dependent upon the frequency offset and/or phase offset (Δf, Δφ).
 2. Method for carrier-frequency synchronisation according to claim 1, wherein the time-discrete phases (2·(2πΔft′+Δφ)) of the complex rotating phasors (|A(t′)|·e^(j2(2πΔf·t′+Δφ))), of the modified vestigial-sideband-modulated received signal (x_(VSB)′(t′)) are dependent only upon the frequency offset and/or phase offset (Δf, Δφ).
 3. Method for carrier-frequency synchronisation according to claim 1, wherein the conversion (S20, S60, S70, S80) of the vestigial-sideband-modulated received signal (r_(VSB) (t)) consists of a down mixing (S20) by a quarter of the symbol frequency $\left( \frac{f_{SVSB}}{4} \right),$ a first sampling (S60) at two sampling values per symbol period (T_(S)), a complex multiplication (S70) and a modulus-scaled squaring (S80).
 4. Method for clock synchronisation according to claim 1, wherein the symbol duration (T_(VSB)) of the vestigial-sideband-modulated received signal (r_(VSB)(t)) is half of the symbol duration (T_(S)) of the received signal (r(t′)).
 5. Method for clock synchronisation according to claim 3, wherein the complex multiplication (S70) takes place with the complex phase angle ${\mathbb{e}}^{{- j}\quad\mu\frac{\pi}{2}},$ wherein μ is the sampling index.
 6. Method for carrier-frequency synchronisation according to claim 3, wherein the modulus-scaled squaring (S80) takes place by parallel squaring, modulus formation and subsequent division.
 7. Method for carrier-frequency synchronisation according to claim 3, wherein in the case of an inverted position of the sideband of the vestigial-sideband-modulated received signal (r_(VSB)(t)), the down mixing (S20) is preceded by a mirroring (S10) of the sideband from the inverted position into the normal position.
 8. Method for carrier-frequency synchronisation according to claim 3, wherein the down mixing (S20) of the vestigial-sideband-modulated received signal (r_(VSB)(t)) is followed by an over-sampling (S30), an estimation filtering (S40) and a clock synchronisation (S50).
 9. Method for carrier-frequency synchronisation according to claim 3, wherein the conversion (S20, S60, S70, S80) of the vestigial-sideband-modulated received signal (r_(VSB)(t)) is followed by a maximum-likelihood estimation (S90) of the frequency offset and phase offset (Δf, Δφ) of the carrier signal (e^(j(2π(f) ^(T) ^(+Δf)t+Δφ))).
 10. Device for carrier-frequency synchronisation of a carrier signal (e^(j(2π(f) ^(T) ^(+Δf)t+Δφ))) affected by a frequency offset and/or phase offset (Δf, Δφ) with a maximum-likelihood estimator (18) for estimating the frequency offset and/or phase offset (Δf, Δφ) of the carrier signal (e^(j(2π(f) ^(T) ^(+Δφ)t+Δφ))) from a received signal (r(t′)) with time-discrete, complex rotating phasors (|r(t′)|·e^(j2πΔf·t′+Δφ)), in which only the timediscrete phases (2πΔft′+Δφ) are dependent upon the frequency offset and/or phase offset (Δf, Δφ), wherein the maximum-likelihood estimator (18) is preceded by a signal-processing unit (15) and a down mixer (11), which converts the received signal (r(t′)) which signal (r(t′)) which is a vestigial-sideband-modulated received signal (r_(VSB)(t)), into a modified vestigial-sideband-modulated received signal (x_(VSB)′(t′)) with time-discrete complex rotating phasors (|A(t′)|·e^(j2(2πΔf·t′+Δφ)), in which only the time-discrete phases (2πΔft′+Δφ) are dependent upon the frequency offset and/or phase offset (Δf, Δφ).
 11. Device for carrier-frequency synchronisation according to claim 10, wherein the signal-processing unit (15) consists of a first sampling unit (12), a complex multiplier (13) and a unit for modulus-scaled squaring (14).
 12. Device for carrier-frequency synchronisation according to claim 11, wherein the unit for modulus-scaled squaring (14) consists of a squaring element and a parallel-connected modulus former with a divider connected downstream of the squaring element and the modulus former.
 13. Device for carrier-frequency synchronisation according to claim 10, wherein the down mixer (11) is preceded by a unit for sideband mirroring (10).
 14. Device for carrier-frequency synchronisation according to claim 10, wherein the down mixer (11) is followed by a second sampling unit (7), an estimation filter (8) and a time-delay unit (9) for clock synchronisation.
 15. Digital storage medium with electronically-readable control signals, which can cooperate in such a manner with a programmable computer or digital signal processor, that the method according to claim 1 is executed.
 16. Computer software product with program-code means stored on a machine-readable carrier, in order to implement all of the stages according to claim 1, when the program is executed on a computer or a digital signal processor.
 17. Computer software with program code means, in order to implement all of the stages according to claim 1, when the program is executed on a computer or a digital signal processor.
 18. Computer software with program-code means, in order to implement all of the stages according to claim 1, when the program is stored on a machine-readable data carrier. 